The analog output of an interferometer represents the combined power of two distinct waves. In general the two waves will interfere resulting in an average power that depends upon the cosine of the phase shift .theta. between the two waves. The analog output V of a detector, which measures the combined power is given by: EQU V=P.sub.dc +P.sub.ac cos(.theta.)
where, P.sub.dc is an offset and P.sub.ac is the gain of the interferometer output. Optical interferometers are the most common devices having this analog output and many utilize fiber optics to guide light waves. In optical sensor applications, an external parameter will proportionally change the phase of one or both of the light waves and the resulting change in the analog output of the light detector of the interferometer is used to sense that parameter. Examples of parameters that can affect the phase of light waves are rotation, pressure and magnetic fields. If R is the input phase shift of the interferometer induced by the parameter and r the phase shift due to other causes, then the analog output is: EQU V=P.sub.dc +P.sub.ac cos(R+r)
This analog output V, must be used in some manner to measure the input phase R to build a useful sensor system. As it normally occurs, the analog output V is not very useful, because it is a nonlinear function of the input phase R, resulting in no output changes and sign ambiguities at many periodic operating points. Also, when R is to be measured with DC accuracy, the offset P.sub.dc obscures the result. Finally, changes in P.sub.dc or P.sub.ac within the bandwidth of R corrupt the measurement.
These well known limitations are the reasons that phase modulation is introduced into the system. One embodiment of the present invention is concerned with sinusoid phase modulations that are commonly known as "Phase Generated Carrier" or PGC approaches from "Homodyne Demodulation Scheme for Fiber Optic Sensors Using Phase Generated Carrier", IEEE Journal of Quantum Electronics, Oct. 1982, QE-18, No 10, pp. 1647-1653; Dandridge et al. In such systems, a device is present in the interferometer that introduces a phase shift at a constant frequency resulting in an analog output of: EQU V(t)=P.sub.dc +P.sub.ac cos{R+M sin(t+W)}
where M is the modulation depth of the interferometer, W is the phase of the modulation of the interferometer and t is the linearly increasing time in units of radians. The modulation phase defined by the term: EQU M sin(t+W)
in the cosine argument is the result of a single frequency sinusoidal drive output applied to the interferometer. The way in which this drive output creates the modulation phase depends on the design of the interferometer. In rotation sensors commonly made out of Sagnac interferometers, the time difference of the sine drive output applied to a phase shifter inside the Sagnac loop gives the modulation. In time domain multiplexed acoustic sensors fabricated from Michelson interferometers, a sinusoidal variation of the light source current will induce wavelength changes which will cause the modulation phase given above. In the simplest modulator case, a fiber wrapped on a piezoelectric cylinder is placed in one arm of a Mach-Zehnder interferometer to transform its sinusoid drive into the modulation phase.
Most approaches to measuring the input phase R in the presence of a sine modulation work with the harmonic series of the analog output given by: ##EQU1## The simplest and most limited open loop interferometric demodulation approach using PGC modulation is where the analog output is mixed with a reference signal equivalent to the frequency used to perform the modulation and low pass filtered with a gain K to the bandwidth of the input phase R. The resulting analog output is: EQU V.sub.1 ={2K P.sub.ac cos(W.sub.1)J.sub.1 (M)}.multidot.sin R
which may be viewed as a scaling factor in the curly brackets multiplied by the sine of the input phase R. When R has an absolute value less than 0.2 radians, the small angle approximation of sin R=R is valid, resulting in a linear demodulated output over the range of .+-.0.2 radians. The terms that make up the scale factor in the brackets point out the possible demodulation errors which this simple approach shares with many other methods. The scale factor will change with variations in the interferometer optical gain Pac, the modulation depth M, the synchronous detection phase W and the filter gain K. In the real world applications, where thermal environments may vary by 100 degrees Celsius or more, it is not uncommon to find any one of these scaling terms to change by 10% or more.
Many of those familiar with the art are knowledgeable of these effects and can, through design processes, provide some mitigation of them. For example, two reference frequencies and two synchronous detection channels will add a quadrature output term as follows: EQU V.sub.2 ={2K P.sub.ac cos(W.sub.2) J.sub.2 (M)}.multidot.cos R
It is evident that, with quadrature measures of the phase, either analog or digital inverse trigonometric post processing may be utilized to determine R. The analog process will be limited in range to .+-..pi./2 radians; where a digital approach can go much further. If the quadrature terms are digitized, it is possible to implement a processing algorithm such as that defined in U.S. Pat. No. 4,789,240 in a process flow detailed in this patent's FIG. 7 (starting at the third step) which provides for an extremely large dynamic range, which is limited only by the constraint that the rate of change of phase not exceed .pi. radians per consecutive sample. This process initially determines R to within a quadrant or octant, and then using the polarity of the quadrature terms, determines R to within the unit circle, and then using previous sample phase information is able to use simple logic to track phase as it crosses fringe boundaries. This tracking capability is limited only to the bit length of the counter used to track fringe crossings. This design and others like it show promise for large dynamic ranges if implemented in a digital format, but its measurement performance still falls short in that it provides no means of correcting scaling error terms related to modulation depth control, synchronous detector phase errors, and cross channel processing gain variations.
It would be a great improvement to the art if the linear dynamic range of an open loop interferometric demodulator could be arbitrarily large (micro-radians to millions of radians or greater) while it implemented processes, which automatically control all critical scaling factors such that they are invariant. Additionally, such a demodulator should be able to work with many different types of two-beam interferometers, such as Sagnac, Mach-Zehnder, Michelson, Low Finesse Fabry Perot, and others, and be able to efficiently demodulate the multiplexed outputs of such sensors positioned in an array.
There are a number of open loop interferometric demodulation designs described or practiced in the art, which intend to overcome the described scaling errors as well as provide larger linear dynamic ranges. U.S. Pat. Nos. 4,704,032 and 4,756,620 describe approaches which provide active compensation for amplitude scaling, but do not compensate the other scaling errors and additionally provide dynamic ranges less than 1 radian. U.S. Pat. Nos. 4,637,722; 4,687,330; 4,707,136; 4,728,192; and 4,779,975 describe approaches and improvements which compensate for, or are immune to amplitude and phase errors. These implementations also extend dynamic range, but are limited to tens of radians or less. U.S. Pat. Nos. 5,202,747; 5,289,259; 5,355,216; and 5,438,411 describe approaches and improvements which are immune to amplitude variations and to some degree, modulation depth and gain scaling variations. However these approaches are still subject to phase errors caused by band limited operation of square-law detectors and their associated electronic amplifiers. Additionally these approaches are limited in dynamic range to less than 100 radians. U.S. Pat. Nos. 4,765,739; 4,776,700; 4,836,676; 5,127,732; and 5,289,257, describe approaches which compensate for, or are immune to modulation depth and amplitude variations and are also capable of operating over large dynamic ranges. However these approaches are still subject to synchronous detection phase offsets and variations (changes in W) and additionally cross-channel gain scaling variations. U.S. Pat. Nos. 4,883,358 and 5,412,472 describe an approach which actively stabilizes field amplitude, modulation depth, and phase errors and is capable of operating over a large dynamic range. However this approach is still subject to cross channel gain scaling variations.
Although a number of the above referenced design techniques approach the desired high accuracy and large dynamic range design objective, the ones which come the closest require complex electronic circuitry. It would also be a great improvement to the art if an interferometric demodulator meeting the desired high accuracy with a large dynamic range be simple in design and implementation, and low cost in manufacture, which are traits inherent in the present invention.